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NCP1027 High-Voltage Switcher for Medium Power Offline SMPS Featuring Low Standby Power
The NCP1027 offers a new solution targeting output power levels from a few watts up to 15 W in a universal mains flyback application. Our proprietary high-voltage technology lets us include a power MOSFET together with a startup current source, all directly connected to the bulk capacitor. To prevent lethal runaway in low input voltage conditions, an adjustable brown-out circuitry blocks the activity until sufficient input level is reached. Current-mode operation together with an adjustable ramp compensation offers superior performance in universal mains applications. Furthermore, an Over Power Protection pin brings the ability to precisely compensate all internal delays in high input voltage conditions and optimize the maximum output current capability. Protection wise, a timer detects an overload or a short-circuit and stops all operations, ensuring a safe auto-recovery, low duty cycle burst operation. An integrated, auto-recovery, Overvoltage Protection permanently monitors the VCC level and temporarily shuts down the driving pulses in case of an unexpected feedback loop runaway. Finally, a great RDS(on) figure makes the circuit an excellent choice for standby/auxiliary offline power supplies or applications requiring higher output power levels.
Features http://onsemi.com MARKING DIAGRAM
8-LEAD PDIP P SUFFIX CASE 626A P1027PXXX AWL YYWWG
XXX = 65 or 100 A = Assembly Location WL = Wafer Lot YY = Year WW = Work Week G = Pb-Free Package
PIN CONNECTIONS
VCC Ramp Comp. Brown-Out FB (Top View) Drain GND OPP
* * * * * * * * * * * * * * *
Built-in 700 V MOSFET with Typical RDS(on) of 5.8 W, TJ = 25C Current-Mode Fixed Frequency Operation: 65 kHz and 100 kHz Fixed Peak Current of 800 mA Skip-Cycle Operation at Low Peak Currents Internal Current Source for Clean and Lossless Startup Sequence Auto-Recovery Output Short Circuit Protection with Timer-Based Detection Auto-Recovery Overvoltage Protection with Auxiliary Winding Operation Programmable Brown-Out Input for Low Input Voltage Detection Programmable Over Power Protection Input to Permanently Latchoff the Part Internal Frequency Jittering for Improved EMI Signature Extended Duty Cycle Operation to 80% Typical No-Load Input Standby Power of 85 mW @ 265 Vac 500 mW Loaded, Input Power of 715 mW @ 230 Vac These are Pb-Free Devices
ORDERING INFORMATION
Device NCP1027P065G NCP1027P100G Package PDIP-8 (Pb-Free) PDIP-8 (Pb-Free) Shipping* 50 Units / Rail 50 Units / Rail
*For additional information on our Pb-Free strategy and soldering details, please download the ON Semiconductor Soldering and Mounting Techniques Reference Manual, SOLDERRM/D.
Typical Applications
* Medium Power AC-DC Adapters for Chargers * Auxiliary/Standby Power Supplies for ATX and TVS Power Supplies
Reference NCP1027 - 5.8 W 230 VAC 25 W* 90-265 VAC 15 W*
*Typical values, open-frame, 65 kHz version, RqJA < 75C/W, TA = 50C.
(c) Semiconductor Components Industries, LLC, 2006
1
December, 2006 - Rev. 3
Publication Order Number: NCP1027/D
NCP1027
Vout + OPP*
NCP1027 OVP + 85-265 VAC BO + + 1 2 3 4 5 8 7
GND Ramp Comp.* *Optional component
Figure 1. Typical Application
PIN FUNCTION DESCRIPTION
Pin No. 1 Symbol VCC Function Powers the Internal Circuitry Ramp Compensation in CCM Brown-Out and Latchoff Input Feedback Signal Input Drain Connection - Over Power Protection The IC Ground Description This pin is connected to an external capacitor of typically 22 mF. The VCC includes an active shunt which serves as an auto-recovery overvoltage protection. To extend the duty cycle operation in Continuous Conduction Mode (CCM), pin 3 offers the ability to inject ramp compensation in the controller. If unused, short this pin to VCC. By monitoring the bulk level via a resistive network, the circuit protects itself from low mains conditions. If an external event brings this pin above 4.0 V, the part fully latches off. By connecting an optocoupler to this pin, the peak current setpoint is adjusted accordingly to the output power demand. The internal drain power switch circuit connection. This unconnected pin ensures adequate creepage distance. Driving this pin reduces the power supply capability in high line conditions. If no Over Power Protection is needed, short this pin to ground. -
2
Ramp Comp.
3
Brown-Out
4 5 - 7 8
FB Drain - OPP GND
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IC1 VCC Vclamp Auto-Recovery OVP + -+ + - + Vlatch 20 ms RC S Q Q R IBO BO VBO + + - VCC < 4 V Reset Timer OPP 50 ms RC VCC Mngt Fault UVLOs 4 V rst VDD GND
LEB
Jittering V Icomp dd
65 kHz or 100 kHz CLOCK S Q Q R
Ip Flag Vcc
Ramp Comp. 25% of lp + - Skip
+
Vdd RFB
-+
Soft-Start
UVLO Drain
FB
+ - Ip Flag
Max Ip Selection Over Power Protection Input
Ramp Compensation
Figure 2. Internal Block Diagram
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MAXIMUM RATINGS
Rating Power Supply Voltage on all Pins, Except Pin 5 (Drain) Drain Voltage Maximum Current into Pin 1 when Activating the 8.7 V Active Clamp Thermal Resistance, Junction-to-Air - PDIP7 Thermal Resistance, Junction-to-Air - PDIP7 with 1.0 cm@ of 35 m Copper Area Maximum Junction Temperature Storage Temperature Range ESD Capability, Human Body Model (HBM) (All Pins Except HV) ESD Capability, Machine Model (MM) Symbol VCC BVdss I_VCC RqJA RqJA TJMAX - - - Value -0.3 to 10 -0.3 to 700 15 100 75 150 -60 to +150 2.0 200 Unit V V mA C/W C/W C C kV V
Stresses exceeding Maximum Ratings may damage the device. Maximum Ratings are stress ratings only. Functional operation above the Recommended Operating Conditions is not implied. Extended exposure to stresses above the Recommended Operating Conditions may affect device reliability. 1. This device series contains ESD protection and exceeds the following tests: Human Body Model 2000 V per Mil-Std-883, Method 3015. Machine Model Method 200 V. 2. This device contains latchup protection and exceeds 100 mA per JEDEC Standard JESD78.
ELECTRICAL CHARACTERISTICS (For typical values TJ = 25C, for min/max values TJ = 0C to +125C, Max TJ = 150C,
VCC = 8.0 V, unless otherwise noted.) Characteristic SUPPLY SECTION AND VCC MANAGEMENT VCC Increasing Level at which the Switcher Starts to Operate VCC Decreasing Level at which the Switcher Stops Operation Hysteresis between VCCON and VCC(min) Offset Voltage above VCCON at which the Internal Clamp Activates VCC Voltage at which the Internal Latch is Reset Internal IC Consumption, MOSFET Switching at POWER SWITCH CIRCUIT Power Switch Circuit On-State Resistance NCP1027 (Id = 100 mA) TJ = 25C TJ = 125C Power Switch Circuit and Startup Breakdown Voltage (ID(off) = 120 mA, TJ = 25C) Power Switch and Startup Breakdown Voltage Off-State Leakage Current TJ = 25C (Vds = 700 V) TJ = 125C (Vds = 700 V) Switching Characteristics (RL = 50 W, Vds Set for Idrain = 0.7 x Ilim) Turn-on Time (90%-10%) Turn-off Time (10%-90%) INTERNAL STARTUP CURRENT SOURCE High-Voltage Current Source, VCC = VCCON - 200 mV High-Voltage Current Source, VCC = 0 VCC Transition Level for IC1 to IC2 Toggling Point 1 1 1 IC1 IC2 VCCTh 3.5 350 - 6.0 650 1.3 8.0 900 - mA mA V 5 RDS(on) - - 5 BVdss Idss(OFF) 5 5 - - 50 30 - - 700 5.8 9.8 - 7.0 11 - V mA W 65 kHz or 100 kHz 1 1 - 1 1 1 VCCON VCC(min) VCChyste VCCclamp VCCreset ICC1 7.9 6.7 - 140 - - 8.5 7.2 1.2 200 4.0 1.4 8.9 7.9 - 300 - 1.9 V V V mV V mA Pin Symbol Min Typ Max Unit
5 5
ton toff
- -
35 35
- -
ns ns
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ELECTRICAL CHARACTERISTICS (continued) (For typical values TJ = 25C, for min/max values TJ = 0C to +125C,
Max TJ = 150C, VCC = 8.0 V, unless otherwise noted.) Characteristic CURRENT COMPARATOR Maximum Internal Current Setpoint, Pin 4 Open, TJ = 25C, FSW = 65 kHz (Note 3) Final Switch Current with a Primary Slope of 200 mA/ms, FSW = 65 kHz (Note 4) Maximum Internal Current Setpoint, Pin 4 Open, TJ = 25C, FSW = 100 kHz (Note 3) Final Switch Current with a Primary Slope of 200 mA/ms, FSW = 100 kHz (Note 4) Setpoint Decrease for a Pin 7 Injected Current of 40 mA, TJ = 25C Voltage Level in Pin 7 at which OPP Starts to Operate Soft-Start Duration Propagation Delay from Current Detection to Drain OFF State Leading Edge Blanking Duration INTERNAL OSCILLATOR Oscillation Frequency (Note 5) 65 kHz Version, TJ = 25C Oscillation Frequency (Note 5) 100 kHz Version, TJ = 25C Frequency Jittering in Percentage of fOSC Jittering Swing Frequency Maximum Duty Cycle FEEDBACK SECTION Internal Pullup Resistor Ramp Compensation Level on Pin 1 - Rramp = 100 kW SKIP CYCLE GENERATION Internal Skip Mode Level, in Percentage of Maximum Peak Current PROTECTIONS Brown-Out Level Brown-Out Hysteresis Current, TJ = 25C (Note 3) Brown-Out Hysteresis Current, TJ = 0C to 125C Fault Validation further to Error Flag Assertion OFF Phase in Fault Mode Latching Voltage on Brown-Out Pin Latch Input Integrating Filter Time Constant OVP Integrating Filter Time Constant VCC Current at which the Switcher Stops Pulsing TEMPERATURE MANAGEMENT Temperature Shutdown Hysteresis in Shutdown - - TSD - 160 - - 40 - - C C 3 3 3 - - 3 3 - 1 VBO IBOhyste IBOhyste TimerON TimerOFF Vlatch TdelBOL TdelOVP IOVP 510 10 - 40 - 3.15 - - 6.0 570 11.5 10 55 440 3.5 20 50 8.5 620 13 - - - 3.85 - - 11 mV mA mA ms ms V ms ms mA - Iskip - 25 - % 4 2 Rupp Rlevel - - 16 2.75 - - kW V - - - - - fOSC 58.5 fOSC 90 fJitter fswing Dmax - - 74 100 "6.0 300 80 110 - - 87 % Hz % 65 71.5 kHz kHz - - - - 7 7 - - - Ipeak_27_CS_ 65 k Ipeak_27_SW_ 65 k Ipeak_27_CS_ 100 k Ipeak_27_SW_ 100 k IOPP IOPPtripV TSS Tprop TLEB 720 - 720 - - - - - - 800 820 800 820 23 1.5 1.0 100 200 880 - 880 - - - - - - mA mA mA mA % V ms ns ns Pin Symbol Min Typ Max Unit
3. See characterization curves for full temperature span evolution. 4. The final switch current is: Ipeak_2X_CS + Tprop x Vin / Lp, with Vin the input voltage and Lp the primary inductor in a flyback. 5. Oscillator frequency is measured with disabled jittering.
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11 8.9 8.8 10 IOVP (mA) 8.7 8.6 VCCON (V) 9 8.5 8.4 8.3 8.2 7 8.1 8.0 6 -20 7.9 0 20 40 60 80 100 TEMPERATURE (C) 120 140 -20 0 20 40 60 80 100 120 140
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TEMPERATURE (C)
Figure 3.
Figure 4.
7.9 7.8 7.7 7.6 VCCMIN (V) VCCClamp (V) 0 20 40 60 80 100 120 140 7.5 7.4 7.3 7.2 7.1 7.0 6.9 6.8 6.7 -20
0.24
0.22 0.20
0.18 0.16
0.14
-20 0 20 40 60 80 100 TEMPERATURE (C) 120 140
TEMPERATURE (C)
Figure 5.
Figure 6.
1.8
900 850 800
1.6 ICC1 (mA) IC2 (mA)
750 700 650 600 550 500 450 400
1.4
1.2
1.0 -20
0
20
40
60
80
100
120
140
350 -20
0
20
40
60
80
100
120
140
TEMPERATURE (C)
TEMPERATURE (C)
Figure 7.
Figure 8.
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8.0 7.5 7.0 6.5 Fosc (kHz) -20 0 20 40 60 80 TEMPERATURE (C) 100 120 140 IC1 (mA) 6.0 5.5 5.0 4.5 4.0 3.5 71.0 70.0 69.0 68.0 67.0 66.0 65.0 64.0 63.0 62.0 61.0 60.0 59.0 -20
0
20
40
60
80
100
120
140
TEMPERATURE (C)
Figure 9.
Figure 10.
120 100 80 Fosc (kHz) 60 40 20 0 -40 -20 Dmax (%) 0 20 40 60 80 100 120 140
87.0 85.0 83.0 81.0 79.0 77.0 75.0
-20
0
20
40
60
80
100
120
140
TEMPERATURE (C)
TEMPERATURE (C)
Figure 11.
Figure 12.
610 600 IBO HYSTERESIS (mA) -20 0 20 40 60 80 100 120 140 590 580 VBO (mV) 570 560 550 540 530 520 510 TEMPERATURE (C)
13.0 12.5 12.0 11.5 11.0 10.5 10.0 9.5 9.0 -20 0 20 40 60 80 100 120 140
TEMPERATURE (C)
Figure 13.
Figure 14.
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880 RDS(on) @ ID = 100 mA (W) 0 20 40 60 80 100 120 140 860 840 Ipeak (mA) 820 800 780 760 740 720 -20 11 10 9 8 7 6 5 4 3 2 -20 0 20 TEMPERATURE (C) 40 60 80 TEMPERATURE (C) 100 120 140
Figure 15.
Figure 16.
3.8 3.7 3.6 Vlatch (V) Iopp (%) 3.5 3.4
30 28 26 24 22 20
3.3 3.2 -20
18 0 20 40 60 80 100 120 140 16 -20
Ipin 7 = 40 mA
0
20
40
60
80
100
120
140
TEMPERATURE (C)
TEMPERATURE (C)
Figure 17.
Figure 18.
400 380 Tleb + Tpropdelay (ns) 360 340 320 300 280 260 240 220 200 -20 0 20 40 60 80 TEMPERATURE (C) 100 120 140 RAMP COMPENSATION LEVEL (V)
2.8
2.7
2.6
2.5 -20
0
20
40
60
80
100
120
140
TEMPERATURE (C)
Figure 19.
Figure 20.
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70 68 66 TimerON (ms) Idss OFF (mA) 0 20 40 60 80 TEMPERATURE (C) 100 120 140 64 62 60 58 56 54 52 50 -20 100 90 80 70 60 50 40 30 20 10 -20 0 20 40 60 80 TEMPERATURE (C) 100 120 140
Figure 21.
Figure 22.
85C 100% 90% IPEAK REDUCTION (%) 80% 70% 60% 50% 40% 30% 20% 10% 0% 0 50 100 150 IOPP (mA) 200 250 -40C 25C -20C 0C 140C 125C
Figure 23. Ipeak Reduction = F(lopp, @ temperature)
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APPLICATION INFORMATION Introduction The NCP1027 offers a complete current-mode control solution and enhances the NCP101X series. The component integrates everything needed to build a rugged and low-cost Switch-Mode Power Supply (SMPS) featuring low standby power. * Current-Mode Operation: The controller uses a current-mode control architecture, which, together with an adjustable ramp compensation circuitry, ensures efficient and stable continuous or discontinuous conduction designs. * 700 V-5.8 W Power Switch Circuit: Due to ON Semiconductor Very High Voltage Integrated Circuit technology, the circuit hosts a high-voltage power switch circuit featuring a 5.8 W RDS(on) - TJ = 25C. This value lets the designer build a 15 W power supply operated on universal mains as long as sufficient copper area exists to lower the junction-to-ambient thermal resistance. An internal current source delivers the startup current, necessary to crank the power supply. * Short-Circuit Protection: By permanently monitoring the feedback line activity, the circuit is able to detect the presence of a short-circuit, immediately reducing the output power for a total system protection. A 55 ms timer is started as soon as the feedback pin asks for the maximum peak current. At the end of this timer, if the fault is still present, then the device enters a safe, auto-recovery burst mode, affected by a fixed 440 ms recurrence. Once the short has disappeared, the controller resumes and goes back to normal operation. The timer duration is fully independent from the VCC capacitor value. * Over Power Protection: A possibility exists to reduce the maximum output power capability in high line conditions. A simple two resistor network wired to the bulk capacitor will program the maximum current reduction for a given input voltage (down to 20% of the maximum peak current).
* Fail-Safe Optocoupler/Overvoltage Protection: As
the auxiliary winding is connected to the VCC pin, an internal active clamp connected between VCC and ground limits the supply dynamics to 8.7 V. In case the current injected in this clamp exceeds a level of 6.0 mA (minimum), the controller immediately stops switching and waits a full timer period before attempting to restart. If the default is gone, the controller resumes operation. If the default is still there, e.g. a broken optocoupler, the controller protects the load through a safe burst mode. Brown-Out Input: A fraction of the input voltage appears on pin 3, due to a resistive divider. If the mains drops below a level adjusted by this resistive divider, the circuit does not switch. As soon as the mains goes back within its normal range, the device resumes operation and operates normally. By adjusting the bridge resistors, it becomes possible to set the brown-out levels (on and off) independently. Latchoff: Pin 3 also welcomes a comparator who offers a way to fully latch the controller. If an external event (e.g. an overtemperature) brings the brown-out pin above 3.5 V, the circuit stays permanently off until the user cycles its VCC down, for instance by unplugging the converter from the mains outlet. Frequency Jittering: The internal clock receives a low frequency modulation which helps smoothing the power supply EMI signature. Soft-Start: A 1.0 ms soft-start ensures a smooth startup sequence, reducing output overshoots. Skip Cycle: If SMPS naturally exhibit a good efficiency at nominal load, they begin to be less efficient when the output power demand diminishes. By skipping unneeded switching cycles, the NCP1027 drastically reduces the power wasted during light load conditions. Experiments carried over the 5.0 V/2.0 A demonstration board reveal a standby power at no-load and 265 Vac of 85 mW and an efficiency for 500 mW output power of 64% at 230 Vac.
*
*
* * *
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NCP1027
Startup Sequence The NCP1027 includes a high-voltage startup circuitry, directly deriving current from the bulk line to charge the
Vbulk
VCC capacitor. Figure 24 details the simplified internal arrangement.
I1 RVCC
+
ICC1 1
IC1
5
I2
Iclamp
-+
+ CVCC Vz = 8.7 V
VCCon VCCoff
+
Iclamp > 6 mA OVP fault
8
Figure 24. Internal Arrangement of the Startup Circuitry
When the power supply is first connected to the mains outlet, the internal current source is biased and charges up the VCC capacitor. When the voltage on this VCC capacitor reaches the VCCON level (typically 8.5 V), the current source turns off, reducing the amount of power being dissipated. At this time, the VCC capacitor only supplies the controller, and the auxiliary supply should take over before VCC collapses below VCC(min). This VCC capacitor, CVCC, must therefore be calculated to hold enough energy so that VCC stays above VCC(min) (7.3 V typical) until the auxiliary voltage fully takes over. An auxiliary winding is needed to maintain the VCC in order to self-supply the switcher. The VCC capacitor has only a supply role and its value does not impact other parameters such as fault duration or the frequency sweep period for instance. As one can see in Figure 24, an internal
active Zener diode, protects the switcher against lethal VCC runaways. This situation can occur if the feedback loop optocoupler fails, for instance, and you would like to protect the converter against an over voltage event. In that case, the internal current increase incurred by the VCC rapid growth triggers the over voltage protection (OVP) circuit and immediately stops the output pulses for 440 ms. Then a new startup attempt takes place to check whether the fault has disappeared or not. The OVP paragraph gives more design details on this particular section. The VCC capacitor can be calculated knowing a) the amount of energy that needs to be stored; b) the time it takes for the auxiliary voltage to appear, and; c) the current consumed by the controller at that time. For a better understanding, Figure 25 shows how the voltage evolves on the VCC capacitor upon startup.
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Figure 25. A typical startup sequence showing the VCC capacitor voltage evolution versus time.
Suppose our power supply takes 10 ms (tstartup) to bring the output voltage to its target value. We know that the switcher consumption is around 2.0 mA (ICC1). Therefore, we can calculate the amount of capacitance we need, to hold VCC above 7.5 V at least for 10 ms while delivering 2.0 mA:
ICC1tstartup or, by replacing with the above values, DVCC C w 2m * 10 m w 20 mF then select a 33 mF for the VCC 1 Cw
capacitor. Fault Condition - Short-Circuit on VCC In some fault situations, a short-circuit can purposely occur between VCC and GND. In high line conditions (VHV = 370 VDC) the current delivered by the startup device will seriously increase the junction temperature. For instance, since IC1 equals 3.0 mA (the min corresponds to the highest TJ), the device would dissipate 370 3m= 1.1 W. To avoid this situation, the controller includes a novel circuitry made of two startup levels, IC1 and IC2. At powerup, as long as VCC is below a 1.3 V level, the source delivers IC1 (around 650 mA typical), then, when VCC reaches 1.3 V, the source smoothly transitions to IC2 and delivers its nominal value. As a result, in case of short-circuit between VCC and GND, the power dissipation will drop to 370 650 m = 240 mW. Figure 26 portrays this particular behavior.
Figure 26. The startup source now features a dual-level startup current.
The first startup period is calculated by the formula C V = I t, which implies a 33 m 1.3/650 m = 66 ms startup time for the first sequence (t1). The second sequence (t2) is obtained by toggling the source to 4.0 mA with a delta V of VCCON - VCCth = 8.5 - 1.5 = 7.0 V, which finally leads to a second startup time of 7 33 m/6.0 m = 39 ms. The total startup time becomes 66 m + 39 m = 105 ms as a typical value. Please note that this calculation is approximated by the presence of the knee in the vicinity of the transition.
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Fault Condition - Output Short-Circuit As soon as VCC reaches VCCON, drive pulses are internally enabled. If everything is correct, the auxiliary winding increases the voltage on the VCC pin as the output voltage rises. During the start-sequence, the controller smoothly ramps up the peak current to Imax setting, e.g. Ipeak_HI, which is reached after a typical period of 1.0 ms. As soon as the peak current setpoint reaches its maximum (during the startup period but also anytime an overload occurs), an internal error flag is asserted, Ipflag, indicating that the system has reached its maximum current limit set point (Ip = Ip max). The assertion of this flag triggers a 55 ms counter. If at counter completion Ipflag remains asserted, all driving pulses are stopped and the part stays off during eight periods of 55 ms (440 ms). A new attempt to restart occurs and will last 55 ms providing the fault is still present. If the fault still affects the output, a safe burst mode is entered, affected by a low duty-cycle operation (11%). When the fault disappears, the power supply quickly resumes operation. Figure 27 depicts this particular mode.
Figure 27. In case of short-circuit or overload, the NCP1027 protects itself and the power supply via a low frequency burst mode. The VCC is maintained by the current source and self-supplies the controller.
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In Figure 27, one can see that the VCC is still alive, testifying for a badly coupled power secondary and primary auxiliary windings. Some situations exist where an output short-circuit make the auxiliary winding collapse before the timer completion. In this particular case, the Undervoltage Lock Out (UVLO) circuitry has the priority and safely cuts off all driving pulses. Figure 28 describes this variation.
Figure 28. The auxiliary winding collapses in presence of a short-circuit. Pulses are immediately stopped as VCC crosses the minimum operating voltage, VCC(min).
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Fault Condition - Output Too Low This particular mode of operation occurs when the feedback is ensured by a two-loop control imposing either constant output voltage (CV) or constant output current (CC), for instance in a battery charger. In CC mode, the output voltage falls down below the original target but the feedback loop is kept closed by the CC controller. For that reason, the controller becomes un-able to detect a real output short-circuit since Ipflag will never be asserted. Due to a good winding coupling, the primary side auxiliary collapsing will ensure a proper fault detection via the UVLO internal circuit. Figure 29 depicts this operating way.
Figure 29. In this particular case, the output goes low but the timer is not started since the FB pin is still held by the optocoupler. Due to the UVLO circuit, the controller safely stops operation at VCC = VCC(min).
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Fault Condition - Low Input Voltage The NCP1027 includes a brown-out circuitry able to protect the power supply in case of low input voltage conditions. Figure 30 shows how internally the NCP1027 monitors the voltage image of the bulk capacitor. Below a given level, the controller blocks the driving pulses, above it, it authorizes them. The internal circuitry, depicted by Figure 30a, offers a way to observe the high-voltage (HV) rail. A resistive divider made of Rupper and Rlower , brings a portion of the HV rail on pin 3. Below the turn-on level, the 10 mA current source IBO is off. Therefore, the turn-on level solely depends on the division ratio brought by the resistive divider.
1
vin
2
vcmp
16.0
160
Vbulk = 100 V
12.0 120
Vbulk = 70 V
Plot1 Vcmp Volts
Vin in Volts
Vbulk Rupper
VDD IBO ON/OFF
8.00
80.0
BO
+ - + VBO
4.00
40.0
BO
Rlower
0
0
2 1
20.0u
60.0u
100u Time in Seconds
140u
180u
Figure 30a. The internal brown-out configuration with an offset current source.
Figure 30b. Simulation results for 100/70 ON/OFF levels.
Figure 30.
To the contrary, when the internal BO signal is high, the IBO source is activated and creates an hysteresis. As a result, it becomes possible to select the turn-on and turn-off levels via a few lines of algebra.
IBO is Off V()) + Vbulk1 IBO is On V()) + Vbulk2 Rlower ) IBO Rlower ) Rupper Rlower Rupper Rlower ) Rupper
(eq. 2)
Rlower Rlower ) Rupper
(eq. 1)
We can now extract Rlower from Equation 1 and plug it into Equation 2, then solve for Rupper :
Rupper + Rlower Rlower + VBO Vbulk1-VBO VBO
Vbulk1-Vbulk2 IBO (Vbulk1-VBO)
If we decide to turn-on our converter for Vbulk1 equals 100 V and turn it off for Vbulk2 equals 70 V, then we obtain: Rupper = 3.0 MW Rlower = 18 kW
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The bridge power dissipation is 3302/3.018 Meg = 36 mW in nominal high-line operation. Figure 30b simulation result confirms our calculations. Figure 31 describes signal variations during a brown-out condition. Please note that output pulses only reappear when VCC reaches VCC(ON), ensuring a clean startup sequence. As in fault mode conditions, the startup source is activated on and off and self-supplies the controller in a Dynamic Self-Supply (DSS) mode.
Figure 31. Signal Evolution During a Brown-Out Condition
Depending on input surge tests, it might be necessary to wire a filtering capacitor between BO and GND (close to the circuit) to avoid adversely triggering the internal latch (unless this is a wanted feature) when the pulse train appears. Latchoff Protection There are some situations where the converter shall be fully turned-off and stay latched. This can happen in the presence of a secondary overvoltage (the feedback loop is
drifting) or when an overtemperature is detected. Secondary monitoring is usually implemented when the coupling between auxiliary and power windings does not lead to a precise primary detection, hence the auto-recovery OVP on pin 1 would not satisfy the precision requirements. Due to the addition of a comparator on the BO pin, a simple external circuit can lift up this pin above VLATCH and permanently disable pulses. The VCC needs to be cycled down below 3.5 V typically to reset the controller.
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VCC Vbulk 20 ms RC
+ -
To permanent latch
Vout
Q1 Rupper
+ Vlatch IBO
VDD
BO NTC Rlower + VBO
+ -
BO
Figure 32. Adding a comparator on the BO pin offers a way to latch-off the controller.
In Figure 32, Q1 is blocked and does not bother the BO measurement as long as the NTC and the optocoupler are not activated. As soon as the secondary optocoupler senses an OVP condition, or the NTC reacts to a high ambient
temperature, Q1 base is brought to ground and the BO pin goes up, permanently latching off the controller. Figure 33 depicts the converter behavior in case of total latch-off.
Figure 33. If the BO pin is lifted up to VLATCH, the controller permanently latches off.
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NCP1027
Auto-Recovery Overvoltage Protection The particular NCP1027 arrangement offers a simple way to prevent output voltage runaway when the optocoupler fails. As Figure 34 shows, an active Zener diode monitors and protects the VCC pin. Below its equivalent breakdown voltage, that is to say 8.7 V typical, no current flows in it. If the auxiliary VCC pushes too much current inside the Zener, then the controller considers an OVP situation and stops the pulses. Figure 34 shows that the insertion of a resistor (Rlimit ) between the auxiliary DC level and the VCC pin is mandatory a) not to damage the internal 8.7 V Zener diode during an overshoot for instance (absolute maximum current is 15 mA) b) to implement the fail-safe optocoupler protection (OVP) as offered by the active clamp. Please note that there cannot be bad interaction between the clamping voltage of the internal Zener and VCCON since this clamping voltage is actually built on top of VCCON with a fixed amount of offset (200 mV typical). Rlimit should be carefully selected to avoid triggering the OVP as we discussed, but also to avoid disturbing the VCC in low/light load conditions. The following details how to evaluate the Rlimit value. Self-supplying controllers in extremely low standby applications often puzzles the designer. Actually, if an SMPS operated at nominal load can deliver an auxiliary voltage of an arbitrary 16 V (Vnom), this voltage can drop below 10 V (Vstby) when entering standby. This is because the recurrence of the switching pulses expands so much, that the low frequency refueling rate of the VCC capacitor is not enough to keep a proper auxiliary voltage. Figure 35 portrays a typical scope shot of an SMPS entering deep standby (output unloaded). Thus, care must be taken when calculating Rlimit 1) to not trigger the VCC overcurrent latch (by injecting 6.0 mA into the active clamp - always use the minimum value for worse case design) in normal operation but 2) not to drop too much voltage over Rlimit when entering standby. Otherwise, the converter will enter burst mode as it will sense an UVLO condition. Based on these recommendations, we are able to bound Rlimit between two equations:
Vstby-VCCON Vnom-Vclamp v Rlimit v ICC1 Itrip
(eq. 3)
Where: Vnom is the auxiliary voltage at nominal load. Vstdby is the auxiliary voltage when standby is entered. Itrip is the current corresponding to the nominal operation. It thus must be selected to avoid false tripping in overshoot conditions. Always use the minimum of the specification for a robust design. ICC1 is the controller consumption. This number slightly decreases compared to ICC1 from the spec since the part in standby does almost not switch. It is around 1.0 mA for the 65 kHz version. VCC(min) is the level above which the auxiliary voltage must be maintained to keep the controller away from the UVLO trip point. It is good to obtain around 8.0 V in order to offer an adequate design margin, e.g. to not reactivate the startup source (which is not a problem in itself if low standby power does not matter).
Drain
VCCON = 8.5 V VCC(min) = 7.5 V + - + Startup Source Rlimit D1
VCC
+ Vclamp = 8.7 V Typ. Latch
+ -
+ + - + I > 6 mA
CVCC
+
CAUX
Laux
Ground
Figure 34. A more detailed view of the NCP1027 offers better insight on how to properly wire an auxiliary winding.
Since Rlimit shall not bother the controller in standby, e.g. keep Vauxiliary to around 8.0 V (as selected above), we purposely select a Vnom well above this value. As explained before, experience shows that a 40% decrease
can be seen on auxiliary windings from nominal operation down to standby mode. Let's select a nominal auxiliary winding of 20 V to offer sufficient margin regarding 8.0 V when in standby (Rlimit also drops voltage in standby...).
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NCP1027
Plugging the values in Equation 3 gives the limits within which Rlimit shall be selected:
20-8.7 v Rlimit v 12-8 , that is to say : 1.8 kW t R limit 6m 1m t 4 kW.
If we design a 65 kHz power supply delivering 12 V, then the ratio between auxiliary and power must be: 12/20 = 0.6. The OVP latch will activate when the clamp current exceeds 6.0 mA. This will occur when Vauxiliary grows up to: 1. 8.7 + 1.8 k (6 m + 1.8 m) 23 V for the first boundary (Rlimit = 1.8 kW). 2. 8.7 + 4 k (6 m + 1.8 m) 40 V for the second boundary (Rlimit = 4.0 kW). Due to a 0.6 ratio between the auxiliary VCC and the power winding, the auxiliary OVP will be seen as a lower overshoot on the real output: 1. 23 0.6 13.8 V 2. 40 0.6 24 V As one can see, tweaking the Rlimit value will allow the selection of a given overvoltage output level. Theoretically predicting the auxiliary drop from nominal to standby is an almost impossible exercise since many parameters are involved, including the converter time constants. Fine tuning of Rlimit thus requires a few iterations and
experiments on a breadboard to check the auxiliary voltage variations but also the output voltage excursion in fault. Once properly adjusted, the fail-safe protection will preclude any lethal voltage runaways in case a problem would occur in the feedback loop.
> 30 ms
Figure 35. The burst frequency becomes so low that it is difficult to keep an adequate level on the auxiliary VCC.
Figure 36 describes the main signal variations when the part operates in auto-recovery OVP.
Figure 36. If the VCC current exceeds a certain threshold, an auto-recovery protection is activated and protects the design.
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NCP1027
Improving the Precision in Auto-Recovery OVP Given the OVP variations the internal trip current dispersion incur, it is sometimes more interesting to explore a different solution, improving the situation to the cost of a minimal amount of surrounding elements. Figure 37 shows that adding a simple Zener diode on top of the limiting resistor, offers a better precision since what matters now is the internal 8.7 V VCC breakdown, plus the Zener voltage. A resistor in series with the Zener diodes keeps the maximum current in the VCC pin below the maximum rating of 15 mA just before tripping the OVP.
VCC Rlimit
D1
+
Laux
Ground
Figure 37. A simple Zener diode added in parallel with Rlimit , allows for a better precision OVP.
Figure 38. Internal logic blocks take a certain amount of time before shutting off the driving pulses in presence of an overcurrent event.
Over Power Compensation Over Power Compensation or Protection (OPP) represents a way to limit the effects of the propagation delay when the converter is supplied from its highest input voltage. The propagation delay naturally extends the power capability of any current-limited converter. Figure 38 explains why. The main parameter is the on slope, that is to say, the pace at which the inductor current grows-up when the power switch closes. For a flyback controller, the slope is given by:
V Son + in Lp
(eq. 4)
At low line, Son is relatively low and does not bother the final peak value. The situation differs at high line and induces a higher peak current. Therefore, the power supply output power capability increases with the input voltage. Let us a take a look at a simple example. Suppose the peak current is 700 mA: Lp = 1.0 mH Vin lowline = 100 Vdc Vin highline = 350 Vdc Ipeak,max = 700 mA tprop = 100 ns
Pout + 1 I2 F Lh 2 peak, final SW p
(eq. 6)
where Lp is the transformer magnetizing/primary inductance and Vin , the input voltage. As the internal logic takes some time to react, the switch gate shutdown does not immediately occur when the maximum power limit is detected (just before activating the overload protection circuit). Clearly speaking, it can take up to 100 ns for the NCP1027 current sense comparator to propagate through the various logical gates before reaching the power switch and finally shutting it off. This is the well-known propagation delay noted tprop . Unfortunately, during this time, the current keeps growing as Figure 38 depicts. The peak current will therefore be troubled by this propagation delay. The formula to obtain the final value is simply:
V Ipeak, final + in t prop ) Ipeak, max Lp
(eq. 5)
Where: Fsw is the switching frequency and h the efficiency. Usually h is bigger in high line conditions than in low line conditions. This formula is valid for a Discontinuous Conduction Mode flyback. From Equation 5, we can calculate the final peak current in both conditions: Ipeak,final = (100/1m) x 100n + 700m = 710 mA at low line. Ipeak,final = (350/1m) x 100n + 700m = 735 mA at high line. From Equation 6, we can have an idea of the maximum output power capability again, in both conditions with respective low and high line efficiency numbers of 78% and 82% for instance: Pout,lowline = 0.5 0.712 1m 65k 0.78 = 12.8 W 2 Pout,highline = 0.5 0.735 1m 65k 0.82 = 14.4 W
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NCP1027
This difference might not be seen as a problem, but some design specifications impose stringent conditions on the maximum output current capability, regardless the line input. Hence the need for an OPP input... Since we want to limit the power to 12.8 W at high line, let us calculate the needed peak current: From equation 6: Ipeak +
2Pout = 693 mA to FSWLph
deliver 12.8 W at high line. Compared to our 735 mA, we need to decrease the setpoint by 6% roughly when Vin equals 350 Vdc. The NCP1027 hosts a special circuitry looking at the couple voltage/current present on pin 7. Figure 39 shows how to arrange components around the controller to obtain Over Power Protection.
Bulk
10 kW and ROPPU made of a series string of 4 1.0 MW resistors plus a 10-turn 1.0 MW potentiometer set at its maximum value. An amp-meter is inserted in series with pin 7 and a volt-meter monitors its voltage with respect to ground. Once the power supply is powered, slowly rotate the potentiometer and observe both voltage and current going up at pin 7. At a certain time, as voltage and current increase, the controller will shut down the power supply. The current at this time is the one we are looking for. Suppose these experiments lead to 80 mA with a pin 7 activation voltage of 2.45 V. Final resistor equations are: VbulkH = 375 Vdc ; the maximum voltage at which OPP must shut down the controller VbulkL = 200 Vdc ; the minimum voltage below which OPP is not activated IOPP = 80 mA ; the current in pin 7 Vf = 2.45 V ; the voltage of pin 7 at the above condition
V -V ROPPL + bulkH bulkL Vf + 27 kW IOPP(VbulkL-Vf) V -V ROPPH + ROPPL bulkL f + 2.2 MW Vf
(eq. 7)
ROPPU Current Setpoint Over Power Protection OPP
(eq. 8)
ROPPL GND
Figure 39. A resistive network reduces the power capability in high-line conditions.
If the OPP feature is not needed for some designs, it is possible to ground it via a copper wire to the adjacent ground pin. This can help to develop a larger copper area in an application where the thermal resistance is an important parameter. Ramp Compensation When operating in Continuous Conduction Mode (CCM), current-mode power supplies can exhibit so-called sub-harmonic oscillations. To cure this problem, the designer must inject ramp compensation. The ramp can either be added to the current sense information or directly subtracted from the feedback signal. Figure 40 details the internal arrangement of the ramp compensation circuitry.
First, you need to know the required injected current and the voltage across pin 7 to start activating OPP. Experiments consist in wiring Figure 39 circuit and running the power supply in conditions where it must shut down (e.g. highest input voltage and maximum output current per specification). For this, ROPPL can be put to
VDD
IRR
Gate Reset
Ramp Vp RR
Control
Figure 40. The Internal Feedback Chain and the Ramp Compensation Network
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NCP1027
The principle consists in selecting the RR resistor, connected from pin 2 to ground, to impose a current IRR in the transistor collector. Np:Ns = 1:N = 1:0.052 transformer turn ratio Lp = 3.8 mH primary inductance We can calculate the off slope, the one actually needed to evaluate Sa , by reflecting the output voltage over the primary inductance. The slope is projected over a complete switching period. Here, we use a 65 kHz part.
) Vf V 6 15u Soff + out TSW + + 455 mA 15 ms 0.052 3.8m NLp
(eq. 10)
Due to the internal sense arrangement, this current slope will become a voltage slope having a value of:
S off + 455m 0.375 + 170 mV 15 ms (eq. 11)
If we chose 50% of this downslope, then the final compensation ramp will present a slope of:
Sa + 170m + 85 mV 15 ms 2
(eq. 12)
We then have:
Figure 41. Maximum Peak Current Setpoint Variations versus Ramp Compensation RR + Vp2.75 k + 2.75 2.75k + 89 kW 85m Sa@TSW
(eq. 13)
The equation to get the right compensation level is the following:
RR + Vp2.75 k Sa@TSW
(eq. 9)
In the above calculations, the internal ESD resistor has purposely been omitted to avoid bringing in another variable. In case no ramp compensation is required, pin 2 must be tied to VCC, the adjacent pin. Soft-Start The NCP1027 features a 1.0 ms soft-start, which reduces the power-on stress, but also contributes to lower the output overshoot. Figure 42 shows a typical operating waveform. The NCP1027 features a novel patented structure which offers a better soft-start ramp, almost ignoring the startup pedestal inherent to traditional current-mode supplies.
where Vp, the total voltage swing, equals 2.75 V. Application example: Suppose we have the following flyback specifications: Vout = 5.0 V output voltage Vf = 1.0 V secondary diode forward drop @ Iout nominal
Figure 42. 1.0 ms Soft-Start Sequence
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NCP1027
Jittering Frequency jittering is a method used to soften the EMI signature by spreading the energy in the vicinity of the main switching component. The NCP1027 offers a "6% deviation of the nominal switching frequency. The sweep
Jitter ramp 68.9kHz
sawtooth is internally generated and modulates the clock up and down with a fixed frequency of 300 Hz. Figure 43 shows the relationship between the jitter ramp and the frequency deviation. It is not possible to externally disable the jitter.
65kHz
61.1kHz adjustable
Internal
sawtooth
Figure 43. Modulation Effects on the Clock Signal by the Jittering Sawtooth
Skip-Cycle Skip cycle offers an efficient way to reduce the standby power by skipping unwanted cycles at light loads. However, the recurrent frequency in skip often enters the audible range and a high peak current obviously generates acoustic noise in the transformer. The noise takes its origins
Nominal peak current
in the resonance of the transformer mechanical structure which is excited by the skipping pulses. A possible solution, successfully implemented in the NCP1200 series, also authorizes skip cycle but only when the power demand as dropped below a given level. This is what Figure 44 shows, as implemented on the NCP1027.
Skip cycle current limit
0
Figure 44. Low Peak Current Skip Cycle Guarantees Noise-Free Operation
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NCP1027
5.0 V/3.0 A Universal Mains Power Supply Due to its low RDS(on), the NCP1027 can be used in universal mains SMPS up to 15 W of continuous power, provided that the chip power dissipation is well under control. That is to say that average power calculations and measurements have been carried and correlated. The design of an SMPS around a monolithic device does not differ from that of a standard circuit using a controller and a MOSFET. However, one needs to be aware of certain characteristics specific of monolithic devices. Let us follow the steps: Vin min = 120 Vdc Vin max = 375 Vdc Vout = 5.0 V Vout = 15 W Operating mode is CCM h = 0.8 1. The lateral MOSFET body-diode shall never be forward biased, either during startup (because of a large leakage inductance) or in normal operation as shown by Figure 45. This condition sets the maximum voltage that can be reflected during toff .
350
250
150
50.0
> 0 !!
50.0
-
1.004M
1.011M
1.018M
1.025M
1.032M
Figure 45. The reflected voltage shall always be greater than the minimum input voltage to avoid the forward biasing of the MOSFET body-diode.
Figure 46. Primary Inductance Current Evolution in CCM
As a result, the Flyback voltage which is reflected on the drain at the switch opening cannot be larger than the input voltage. When selecting components, you thus must adopt a turn ratio which adheres to the following equation: N(Vout ) Vf) t Vin, min t Vin min (eq. 14) . In our case, since we operate from a 120 V DC rail while delivering 5.0 V, we can select a reflected voltage of 110 V DC maximum: 120-110 > 0. Therefore, the turn ratio Np:Ns must be smaller than
Np : Ns t 19. We will see later on how it affects the calculation. 2. Lateral MOSFETs have a poorly doped body-diode which naturally limits their ability to sustain the avalanche. A traditional RCD clamping network shall thus be installed to protect the MOSFET. In some low power applications, a simple capacitor can also be used since Vdrain max + Vin ) N (Vout ) Vf) Lf (eq. 15) , where Lf is the leakage Ctot inductance, Ctot the total capacitance at the drain ) Ipeak Vin + 110 + 18.3 or 5)1 Vout ) Vf
turn ratio, Vout the output voltage, Vf the secondary diode forward drop and finally, Ipeak the maximum peak current. Worse case occurs when the SMPS is very close to regulation, e.g. the Vout target is almost reached and Ipeak is still pushed to the maximum. For this design, we have selected our maximum voltage around 650 V (at Vin = 375 Vdc). This voltage is given by the RCD clamp installed from the drain to the bulk voltage. We will see how to calculate it later on. 3. Calculate the maximum operating duty-cycle for this flyback converter operated in CCM:
d max + NVout 1 + + 0.49 Vin,min NVout ) Vin, min 1 ) NV
out
(eq. 16)
4. To obtain the primary inductance, we have the choice between two equations:
L+ (Vind)2 fSWKPin
(eq. 17)
, where K +
DIL and I1
node (which is increased by the capacitor you will wire between drain and source), N the Np:Ns
*
defines the amount of ripple we want in CCM (see Figure 46). Small K: deep CCM, implying a large primary inductance, a low bandwidth and a large leakage inductance.
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NCP1027 * Large K: approaching BCM where the rms losses are
the worse, but smaller inductance, leading to a better leakage inductance. From Equation 16, a K factor of 0.8 (40% ripple), gives an inductance of:
(120 0.49)2 L+ + 3.8 mH 60k 0.8 18.75 Vd DIL + in + 120 3.8m LFSW 0.49 + 258 mA peak * to * peak 60k
If we take the maximum RDS(on) for a 120C junction temperature, i.e. 11 W, then conduction losses worse case are:
Pcond + I2d, rms Rds(on) + 571 mW
6. Off-time and on-time switching losses can be estimated based on the following calculations:
Poff + IpeakVdstoff + 0.447 650 40n + 130 mW 6 15u 6TSW Pon + IpeakN(Vout ) Vf)ton 6TSW + 0.447 114 40n + 22 mW 6 15u
(eq. 18)
The peak current can be evaluated to be:
Iavg DIL DI Ipeak + ) + Ipeak + 156m ) L + 447 mA 0.49 2 2 d
(eq. 19)
In Figure 46, I1 can also be calculated:
DI I1 + Ipeak- L + 0.447-0.129 + 318 mA 2
5. Based on the above numbers, we can now evaluate the conduction losses:
Id, rms + I1 d 1 ) 1 DIL 2 + 0.318 3 2I1 0.258 2 + 228 mA rms 1)1 3 2 0.318 0.7
The theoretical total power is then 0.571 + 0.13 + 0.022 = 723 mW. 7. The ramp compensation will be calculated as suggested by Equation 13 giving a resistor of 78 kW or 82 kW for the normalized value. Power Switch Circuit Protection As in any Flyback design, it is important to limit the drain excursion to a safe value, e.g. below the power switch circuit BVdss which is 700 V. Figures 47a, b, c present possible implementations:
HV Rclamp
HV
HV
Cclamp D
Dz D
1 2 3 CVCC + 4
8 7 6 5 CVCC + C
1 2 3 4
8 7 6 5 CVCC +
1 2 3 4
8 7 6 5
a.
b.
c.
Figure 47. Different Options to Clamp the Leakage Spike
Figure 47a: The simple capacitor limits the voltage according to Equation 14. This option is only valid for low power applications, e.g. below 5.0 W, otherwise chances exist to destroy the MOSFET. After evaluating the leakage inductance, you can compute C with Equation 15. Typical values are between 100 pF and up to 470 pF. Large capacitors increase capacitive losses...
Figure 47b: The most standard circuitry called the RCD network. You calculate Rclamp and Cclamp using the following formulae:
Rclamp + 2Vclamp(Vclamp-(Vout ) Vf) N) LpeakI2 peak FSW
(eq. 20)
Vclamp Cclamp + VrippleFSWRclamp
(eq. 21)
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NCP1027
clamp is usually selected 50-80 V above the reflected value N (Vout + Vf). The diode needs to be a fast one and an MUR160 represents a good choice. One major drawback of the RCD network lies in its dependency upon the peak current. Worse case occurs when Ipeak and Vin are maximum and Vout is close to reach the steady-state value. Figure 47c: This option is probably the most expensive of all three but it offers the best protection degree. If you need a very precise clamping level, you must implement a Zener diode or a TVS. There are little technology differences behind a standard Zener diode and a TVS. However, the die area is far bigger for a transient suppressor than that of Zener. A 5.0 W Zener diode, like the 1N5388B, will accept 180 W peak power if it lasts less than 8.3 ms. If the peak current in the worse case (e.g. when the PWM circuit maximum current limit works) multiplied by the nominal Zener voltage exceeds these 180 W, then the diode will be destroyed when the supply experiences overloads. A transient suppressor like the P6KE200 still dissipates 5.0 W of continuous power, but is able to accept surges up to 600 W @ 1.0 ms. Select the Zener or TVS clamping level between 40 to 80 V above the reflected output voltage when the supply is heavily loaded.
Power Dissipation and Heatsinking The NCP1027 hosting a power switch circuit and a controller, it is mandatory to properly manage the heat generated by losses. If no precaution is taken, risks exist to trigger the internal thermal shutdown (TSD). To help dissipating the heat, the PCB designer must foresee large copper areas around the PDI7 package. When surrounded by a surface greater than 1.0 cm@ of 35 mm copper, it becomes possible to drop the thermal resistance junction-to-ambient, RqJA down to 75C/W and thus dissipate more power. The maximum power the device can thus evacuate is: P max +
Tj max -Tamb max (eq. 22) RqJA
which gives around 930 mW for an ambient of 50C and a maximum junction of 120C. The losses inherent to the switch circuit RDS(on) can be theoretically evaluated, but the final prototype evaluation must include board measurements to confirm that the junction temperature stays within safe limits. Figure 48 gives a possible layout to help dropping the thermal resistance. When measured on a 70 mm (2 oz.) copper thickness PCB, we obtained a thermal resistance of 75C/W.
Figure 48. A possible PCB arrangement to reduce the thermal resistance junction-to-ambient.
When routing the printed circuit, it is important to keep high impedance line very short, like the brown-out signal and the OPP input if used. Application Diagram Figure 49 displays the final application schematic. The output uses a TLV431 whose low bias current represents an advantage for low standby power switch mode supplies. The secondary side features an additional LC filter needed
to remove unwanted spikes, although less problematic than in DCM operation. On the primary side, a resistive network senses the input bulk voltage and prevents the controller from turning on for input voltages below 100 Vdc. The auxiliary winding delivers 20 V nominal and thus offers comfortable margin when the converter enters standby. As we do not use any OPP, pin 7 goes to ground and offers extended possibility to layout more copper area.
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Np:Ns = 1:0.062 Np:Naux = 1:0.208 Lp = 3.8 mH C5 C8 C9 D1 470 470 470 MBRD640CTT4 mF mF mF + + + R4 100 D5 1N4637
L2 2.2 mH
Vout 5V@ 3A
R5 150 k Type = 1W D2 1N4637
C7 10 nF Type = 400 V
R11 1k
R6 10 k
R1 2.8 M 20 V + C11 1 mF R9 5.6 k 1 2 3 + R3 18 k CVCC 47 mF C1 R2 10 n 78 k 4 C12 100 p 5 U2 NCP1027 8 7
+ C3 100 mF
R10 200 k + C4 47 mF/ 400 V
NCP1027
Figure 49. 5.0 V-3.0 A Universal Mains Power Supply
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C13 220 nF 85-265 VAC Type = X2
U1 TL431
C2 100 nF
R7 10 k
C10 2.2 nF Type = Y1
NCP1027
Transformer Specifications: Vout = 5.0 V/3.0 A Vaux = 20 V/10 mA Lp = 3.8 mH Ip, rms = 280 mA Ip, max = 800 mA Isec, rms = 5.0 A Fsw = 65 kHz Np:Nsec = 1 : 0.052 Np:Naux = 1 : 0.208
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NCP1027
PACKAGE DIMENSIONS
8-LEAD PDIP P SUFFIX CASE 626A-01 ISSUE O
8
5
B
1 4
L M J
NOTES: 1. DIMENSIONING AND TOLERANCING PER ANSI Y14.5M, 1982. 2. CONTROLLING DIMENSION: MILLIMETER. 3. PACKAGE CONTOUR OPTIONAL (ROUND OR SQUARE CORNERS). 4. DIMENSION L TO CENTER OF LEAD WHEN FORMED PARALLEL. 5. DIMENSIONS A AND B ARE DATUMS. DIM A B C D F G H J K L M N MILLIMETERS MIN MAX 9.40 10.16 6.10 6.60 3.94 4.45 0.38 0.51 1.02 1.78 2.54 BSC 0.76 1.27 0.20 0.30 2.92 3.43 7.62 BSC --- 10_ 0.76 1.01 INCHES MIN MAX 0.370 0.400 0.240 0.260 0.155 0.175 0.015 0.020 0.040 0.070 0.100 BSC 0.030 0.050 0.008 0.012 0.115 0.135 0.300 BSC --- 10_ 0.030 0.040
F
NOTE 3
A
C -T-
SEATING PLANE
N D K
M
H
G 0.13 (0.005) TA
M
B
M
The product described herein (NCP1027), may be covered by one or more of the following U.S. patents: 5,418,410; 5,477,175; 6,271,735; 6,362,067; 6,385,060; 6,429,709; 6,448,625; 6,492,679; 6,597,221; 6,633,193; 6,919,598; 6,940,320. There may be other patents pending.
ON Semiconductor and are registered trademarks of Semiconductor Components Industries, LLC (SCILLC). SCILLC reserves the right to make changes without further notice to any products herein. SCILLC makes no warranty, representation or guarantee regarding the suitability of its products for any particular purpose, nor does SCILLC assume any liability arising out of the application or use of any product or circuit, and specifically disclaims any and all liability, including without limitation special, consequential or incidental damages. "Typical" parameters which may be provided in SCILLC data sheets and/or specifications can and do vary in different applications and actual performance may vary over time. All operating parameters, including "Typicals" must be validated for each customer application by customer's technical experts. SCILLC does not convey any license under its patent rights nor the rights of others. SCILLC products are not designed, intended, or authorized for use as components in systems intended for surgical implant into the body, or other applications intended to support or sustain life, or for any other application in which the failure of the SCILLC product could create a situation where personal injury or death may occur. Should Buyer purchase or use SCILLC products for any such unintended or unauthorized application, Buyer shall indemnify and hold SCILLC and its officers, employees, subsidiaries, affiliates, and distributors harmless against all claims, costs, damages, and expenses, and reasonable attorney fees arising out of, directly or indirectly, any claim of personal injury or death associated with such unintended or unauthorized use, even if such claim alleges that SCILLC was negligent regarding the design or manufacture of the part. SCILLC is an Equal Opportunity/Affirmative Action Employer. This literature is subject to all applicable copyright laws and is not for resale in any manner.
PUBLICATION ORDERING INFORMATION
LITERATURE FULFILLMENT: Literature Distribution Center for ON Semiconductor P.O. Box 5163, Denver, Colorado 80217 USA Phone: 303-675-2175 or 800-344-3860 Toll Free USA/Canada Fax: 303-675-2176 or 800-344-3867 Toll Free USA/Canada Email: orderlit@onsemi.com N. American Technical Support: 800-282-9855 Toll Free USA/Canada Europe, Middle East and Africa Technical Support: Phone: 421 33 790 2910 Japan Customer Focus Center Phone: 81-3-5773-3850 ON Semiconductor Website: www.onsemi.com Order Literature: http://www.onsemi.com/orderlit For additional information, please contact your local Sales Representative
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NCP1027D

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